Strobe flash lamp power supply with input voltage feedthrough afterflow prevention circuit

ABSTRACT

The DC to DC converter includes a coupled inductor having a primary winding and a feedback winding. A drive current regulator circuit receives a variable input voltage from the feedback winding but transmits a constant base drive current to the base terminal of the converter switching transistor. The base drive regulator circuit thereby enables the DC to DC converter to operate at high levels of efficiency over wide ranges of DC input voltages such as twelve to forty-eight volts DC.

This is a continuation of application Ser. No. 42,357, filed 04/24/87now U.S. Pat. No. 4,775,821, which is a continuation-in-part of allowedU.S. application Ser. No. 794,415, filed on 11/4/85 and now U.S. Pat.No. 4,755,723.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to power supplies, and more particularly to DC toDC converter circuits commonly utilized to energize gaseous dischargetubes in the form of strobe flash lamps. The disclosure of that allowedpatent application is hereby incorporated by reference.

2. Description of the Prior Art

Prior art DC to DC converter circuits of the type utilized to energizestrobe flash lamps are typically designed to operate at a single inputvoltage. For vehicular applications, such power supplies are typicallydesigned to operate at either one of the following input voltage levels:twelve volts DC, twenty-four volts DC, thirty-six volts DC orforty-eight volts DC. A wholesaler or distributor who suppliesvehicular-mounted strobe power supplies must normally carry an inventoryof strobe power supply circuits for each of these four distinct andpreviously incompatible power supply input voltage ratings.

Strobe power supplies for vehicular applications typically include DC toDC converter circuits using a coupled inductor having primary andfeedback windings which are controlled by a switching transistor coupledin series with the current flow path of the primary winding. In allcases, it is necessary to miniaturize such power supply circuits to themaximum extent possible to render them compatible with vehicularinstallations. In addition, the efficiency of these power supplycircuits must be as high as possible due to the relatively limitedelectrical power generating capacity of the vehicles on which such powersupply circuits are mounted. As is always the case, the cost of suchcircuits must be kept as low as possible and the reliability of thesecircuits which are exposed to extreme environmental conditions must beas high as possible.

From the customer and distributor standpoint, this prior art solution tothe numerous problems addressed above has been begrudgingly accepted,but the additional cost and inconvenience caused by the requirement forstocking and using four different input voltage capacity power supplieshas long been recognized.

SUMMARY OF THE INVENTION

It is therefore a primary object of the present invention to provide adrive current regulator circuit for a variable input voltage DC to DCconverter for supplying a constant base drive current to the convertercircuit switching transistor which does not vary in response tosubstantial changes in the converter DC input voltage.

Another object of the present invention is to provide a drive currentregulator circuit for a variable input voltage DC to DC converter whichcouples a constant current source to the feedback winding of theconverter coupled inductor to provide a constant base drive current tothe converter switching transistor regardless of feedback windingvoltage or the converter DC input voltage.

Another object of the present invention is to provide a drive currentregulator circuit for a variable input voltage DC to DC converter whichcan provide a constant, optimum level base drive current to theconverter switching transistor to maintain optimum, highest efficiencyperformance of the converter circuit where circuit performance isessentially insensitive to substantial variations in the converter DCinput voltage.

Another object of the present invention is to provide an afterglowprevention circuit for a DC to DC converter which prevents strobe flashlamp afterglow caused by the transfer of high level converter DC inputvoltages to the flash lamp during the time interval immediately afterthe converter energy storage capacitor has completed its discharge cyclethrough the strobe flash lamp to thereby prevent afterglow.

Another object of the present invention is to provide an afterglowprevention circuit for a DC to DC converter which utilizes a variableimpedance device in the form of a capacitor placed in series between theconverter switching transistor and the converter energy storagecapacitor to provide a low impedance path between the primary winding ofthe converter coupled inductor and the energy storage capacitor when theconverter operates at high frequencies and to provide a high impedancepath between the primary winding of the coupled inductor and the energystorage capacitor when the converter operates at low frequencies.

Another object of the present invention is to provide an afterglowprevention circuit for a DC to DC converter which can readily be adaptedto function in connection with prior art DC to DC converter circuits tototally eliminate afterglow problems caused by power supply inputvoltage feed through to an ionized strobe flash lamp at the end of theenergy storage capacitor discharge cycle.

Another object of the present invention is to provide an overvoltageprotection circuit for a DC to DC converter which limits the maximumconverter output voltage without affecting the operating efficiency ofthe converter circuit.

Another object of the present invention is to provide an overvoltageprotection circuit for a DC to DC converter which can be added to priorart DC to DC converter circuits without modification of the existingcircuitry and with the addition of only a limited number of additionalcircuit elements.

Another object of the present invention is to provide an overvoltageprotection circuit for a DC to DC converter which utilizes asemiconductor device in the form of a sidac having a breakover voltageand a negative resistance region.

Another object of the present invention is to provide an overvoltageprotection circuit for a DC to DC converter which completely terminatesthe operation of the converter switching transistor for a predeterminedtime period enabling the energy storage capacitor output voltage to bedecreased to a level below the maximum desired output voltage before theconverter switching transistor is once again enabled to commence normaloperation.

Briefly stated, and in accord with one embodiment of the invention, a DCto DC converter circuit receives an input voltage residing within arange of between a minimum input voltage and a maximum input voltage anddelivers current to an intermittently energized load. The emitter andcollector terminals of a switching transistor are coupled in series withthe primary winding of the coupled inductor. The switching transistortransitions between conductive and non-conductive states to control theflow of current through the coupled inductor primary winding. A driveregulator circuit is coupled between the feedback winding of the coupledinductor and the base terminal of the switching transistor to controlthe flow of current from the feedback winding into the switchingtransistor base to maintain the base current flow fixed within a rangedefined by a predetermined maximum current level and a predeterminedminimum current level, where the minimum current level is adequate tomaintain the switching transistor in the conductive state. The driveregulator circuit operates as stated for any input voltage level lyingbetween the minimum and maximum input voltage levels. A load is coupledto the primary winding of the coupled inductor and receives energystored in the coupled inductor after the switching transistor has beenswitched into the non-conductive state. Regardless of extraordinarilywide variations in the converter input voltage, the drive regulatorcircuit of the present invention is able to maintain and direct anoptimal level of base drive current to the switching transistor toprovide safe and highly efficient operation of the switching transistorand related DC to DC converter circuit elements regardless of suchvariations in the converter DC input voltage.

DESCRIPTION OF THE DRAWINGS

The invention is pointed out with particularly in the appended claims.However, other objects and advantages together with the operation of theinvention may be better understood by reference to the followingdetailed description taken in connection with the followingillustrations, wherein:

FIG. 1 is a schematic diagram of a first preferred embodiment of theinvention including the drive current regulator circuit, the afterglowprevention circuit and the overvoltage protection circuit aspects of thepresent invention.

FIG. 2 is a partial schematic diagram depicting an alternative preferredembodiment of the afterglow prevention circuit of the present inventionwhich operates in combination with the overvoltage protection circuit ofthe present invention to maintain a constant magnitude energy dischargeinto the strobe flash lamp regardless of variations in the DC to DCconverter circuit input voltage.

FIG. 3 is a timing diagram showing the relationship to time of theswitching transistor operating frequency with respect to the energystorage capacitor voltage as well as the charge and discharge cycles ofthe converter energy storage capacitor.

FIG. 4 is a partial schematic diagram of the FIG. 1 DC to DC convertercircuit particularly illustrating the manner in which the variableimpedance means of the present invention blocks current flow caused bythe converter DC input voltage into the strobe flash lamp when it is inthe ionized state.

FIGS. 5A, 5B and 5C represent voltage versus current plots for negativeresistance devices in the form of a sidac, a diac and a siliconbilateral switch which are capable of functioning in the overvoltageprotection circuit of the present invention.

FIG. 6 represents a partial schematic diagram of a prior art resistivebiasing circuit for a DC to DC converter.

FIGS. 7A graphically illustrates variations in feedback winding voltagewith respect to variations in input voltage. FIG. 7B illustratesvariations in the FIG. 6 switching transistor base current in responseto variations in the converter input voltage. FIG. 7C illustrates theabsence of variations of the switching transistor base drive currentwith respect to input voltage variations for the drive current regulatorcircuit of the present invention.

FIG. 8 is a partial schematic diagram illustrating the use of a constantcurrent diode in the drive regulator circuit of the present invention inplace of the more complicated constant current source drive regulatorcircuit depicted in the FIG. 1 schematic diagram.

DESCRIPTION OF THE PREFERRED EMBODIMENT

In order to better illustrate the advantages of the invention and itscontributions to the art, a preferred hardware embodiment of theinvention will now be described in detail.

Referring now to FIG. 1, the DC to DC converter of the present inventionincludes a coupled inductor having a primary winding 10 and a feedbackwinding 12. The collector terminal of a switching transistor 14 iscoupled to primary winding 10 while the emitter of that transistor iscoupled to ground through emitter load resistor 16.

An input voltage residing generally between the range of twelve toforty-eight volts DC is coupled across voltage input terminal 18 andground terminal 20. A starting base bias current is provided to the baseterminal of switching transistor 14 by current flow through resistor 22.This starting bias current turns switching transistor 14 on and causesthe current through transistor 14 to increase. When the current levelthrough switching transistor 14 reaches a predetermined maximum value,the voltage on emitter resistor 16 activates power supply disablingmeans which includes transistor 24 and diode 26. The value of emitterresistor 16 is selected so that diode 26 commences conducting current atexactly the point when the collector current of switching transistor 14reaches the maximum desired value. The flow of current through diode 26indicated by current flow arrow 28 turns on transistor 24 and shunts allbase drive current to ground as indicated by current flow arrow 30.

As current initially begins to flow through primary winding 10 andswitching transistor 14, a positive voltage is generated across feedbackwinding 12 as indicated by the "+" sign depicted in FIG. 1. Thispositive voltage is coupled to the input terminal of drive regulatormeans 32 which includes a constant voltage biasing network includingzener diode 34 and resistor 36. The configuration of the coupledinductor including primary winding 10 and feedback winding 12, the turnsratios of these windings and related transformer design parameters areconfigured so that the positive voltage generated by feedback winding 12immediately causes zener diode 34 to break down and begin conductingcurrent as indicated by current flow arrow 38. This constant voltagebiasing network produces a constant bias voltage which does not vary inresponse to changes in the converter input voltage within the designparameters of twelve to forty-eight volts DC. This constant bias voltagepoint is directly coupled to the base of constant current transistor 40.Transistor 40 immediately begins generating a constant current outputwhich is directed through diode 42 and into the base terminal ofswitching transistor 14 as indicated by current flow arrow 44. Thearrival of this base drive current on the base terminal of switchingtransistor 14 maintains transistor 14 in the "on" or conductive stateuntil transistor 24 of the power supply disabling means is once againactivated to shunt the base drive current to ground in response to thebias voltage produced across emitter resistor 16.

The magnitude of the base drive current produced by drive regulatormeans 32 can be adjusted between minimum and maximum allowable levels orto an optimum base drive level by adjusting potentiometer 46.

Referring now to FIG. 7, FIG. 7A illustrates the changes in feedbackwinding voltage produced in response to changes in the converter inputvoltage and illustrates the manner in which the feedback winding voltageincreases in response to increases in the converter input voltage.

FIG. 7B illustrates the manner in which the base drive current 44generated by the FIG. 6 prior art resistive drive regulator circuitvaries linearly in response to changes in the converter input voltage.For a comparatively low input voltage such as twelve volts, FIG. 7Bindicates that the base drive current produced by the prior art FIG. 6circuit fails to meet the minimum required base drive requirements ofswitching transistor 14 and would therefore fail to properly operateswitching transistor 14. For intermediate level input voltages on theorder of twenty-four to thirty-six volts, the base drive currentprovided by the FIG. 6 prior art circuit varies either below or abovethe desired optimum base drive current. For higher levels of inputvoltage on the order of about forty-eight volts DC, FIG. 7B illustratesthat the FIG. 6 prior art base drive circuit produces levels of basedrive current in excess of the maximum allowable drive level. Excessbase drive can cause excessive switching transistor current loads,excessive switching transistor temperatures due to extended transistorstorage time and typically results in damage or failure of varioussystem components.

A significant additional problem experienced by the FIG. 6 prior artbase drive circuit relates to the power dissipated by base driveresistor 48. With a four fold increase in converter input voltage fromtwelve to forty-eight volts, the current through resistor 48 increasesfour fold, but the power dissipated by resistor 48 increases sixteenfold. This creates serious problems relating to requirements forutilizing large, high power dissipation resistors and general circuitheat dissipation problems as well as reduced circuit efficiencyresulting from the excessive power dissipation in resistor 48. Inaddition, as explained above in connection with FIG. 7B, the FIG. 6resistive base drive regulation circuit is not capable of maintaining anoptimum base drive current as the converter input voltage varies.

As indicated by the FIG. 7C plot of input voltage versus base drivecurrent variations achieved by the drive regulator means 32 depicted inFIG. 1, the FIG. 1 circuit is able to provide an absolutely constantlevel of base drive current regardless of fluctuations in converterinput voltage. Appropriate adjustments in potentiometer 46 or anappropriate selection of a fixed resistor value in place ofpotentiometer 46 can readily cause the FIG. 1 drive regulator means 32to produce and maintain a constant base drive current for switchingtransistor 14 to cause that transistor to operate at optimum efficiencyregardless of fluctuations of converter input voltage between the designvalues of from twelve to forty-eight volts DC. With appropriatemodifications the circuit element parameters, drive regulator means 32can readily be adapted to function for various different converter inputvoltage ranges in a manner well known to those of ordinary skill in theart.

Referring now to FIG. 8, an alternative embodiment of drive regulatormeans 32 is illustrated in which a constant current diode 50 has beensubstituted for the related circuit elements depicted in FIG. 1. Withsome limitations, constant current diode 50 has the capability offunctioning to maintain the base drive current constant regardless offluctuations in the converter input voltage.

Referring now to FIGS. 1 and 4, when switching transistor 14 is switchedinto the non-conductive state by base drive shunt transistor 24, currentflows from primary winding 10 of the coupled inductor through variableimpedance means 52 which typically assumes the form of capacitor 52.This current then flows through diode 54 into energy storage means orenergy storage capacitor 56. This allowed charging current flow definesa second current flow path illustrated in FIGS. 1 and 4 by current flowlines 58.

According to well known transformer operating principles as applied tothe converter circuit depicted in FIG. 1, a constant amount of energy istransferred from transformer 10 into capacitor 56 during each timeinterval that transistor 14 is maintained in the non-conductive state.Referring now also to FIG. 3, when the voltage on capacitor 56 is low asis the case at the beginning of the capacitor charge cycle, acomparatively long period of time is required to transfer a fixed amountof energy or energy from primary winding 10 into capacitor 56. Over aperiod of charging cycles, the voltage level of capacitor 56 graduallyincreases as depicted in FIG. 3 until capacitor 56 reaches its peakvoltage at the end of a given charging cycle. At this particular pointduring the charging cycle, only a comparatively short time is requiredto transfer the fixed amount of energy from primary winding 10 intocapacitor 56. The net result of this significant decrease in the timethat switching transistor is maintained in the non-conductive state isthat the operating frequency of switching transistor 14 varies from acomparatively low frequency on the order of about 5,000 to 10,000 Hertzto a comparatively high frequency on the order of about 40,000 to 50,000Hertz for the preferred embodiment of the invention depicted in FIG. 1.

Variable impedance means or capacitor 52 is selected to have acomparatively high level impedance at the beginning of a charging cyclewhich as indicated in FIG. 3 corresponds directly with the end of theflash interval. Similarly, because of the substantial variation inoperating frequency between the beginning and end of any charge cycle,variable impedance means or capacitor 52 can easily be selected to havea substantially lower impedance at the high operating frequency end ofthe charge cycle than was the case with its comparatively low impedancelevel at the beginning of the charge cycle. Because the voltage oncapacitor 56 increases extremely rapidly at the beginning of the chargecycle, any adverse effect of the comparatively high impedance level ofcapacitor 52 at the beginning of the charge cycle is quickly eliminated.

Variable impedance means 52 was positioned as illustrated in FIGS. 1 and2 and selected to have the impedance levels described above to preventafterglow of flash lamp 60.

With prior art converter circuits, input voltages pass through primarywinding 10 and are directly coupled to terminal 62 of flash lamp 60regardless of the operating state of switching transistor 14. When thegaseous interior of flash lamp 60 is initially ionized by triggercircuit 64, the effective impedance of flash lamp 60 drops to anextremely low value on the order of a few Ohms. As illustrated by FIG.3, upon initial ionization of flash lamp 60 by trigger circuit 64,energy storage means 56 rapidly discharges through flash lamp 60 to alow voltage level determined by the holding current of flash lamp 60.With high level converter input voltages on the order of forty-eightvolts DC coupled directly to flash lamp 60, that input voltage by itselfis able to maintain flash lamp 60 in the ionized state and createsafterglow illumination during the time interval illustrated in FIG. 3.

With the FIG. 1 circuit, variable impedance means 52 totally blocks theflow of DC current between converter input terminal 18 and flash lamp 60and totally eliminates the afterglow problem common in prior art unitsas discussed above.

Referring now to FIG. 4, the dotted line identified by reference number66 illustrate a first current flow path and represents the blockedcurrent flow from voltage input terminal 18 to flash lamp 60 achieved byvariable impedance means 52. Without such current blocking, high inputvoltages on terminal 18 which would cause the undesirable afterglowproblem described above. In addition, energy storage means 56 ispermitted to discharge in a normal manner through flash lamp 60 withoutany interference from variable impedance means 52 to thereby form athird current flow path represented by dotted line 68. Because variableimpedance means 52 essentially decouples the potentially high levelconverter input voltage from flash lamp 60, the termination of currentdischarge from energy storage means 56 causes the gaseous interior offlash lamp 60 to rapidly deionize which instantaneously terminates flashlamp illumination. As illustrated in FIG. 3, the converter immediatelyresumes its next capacitor charging cycle at that point in time.

Another design problem encountered and solved by the FIG. 1 circuitrelates to problems arising from either burnout or removal of flash lamp60. Either condition renders flash lamp 60 essentially invisible in theFIG. 1 circuit and the output voltage generated on output voltageconductor 70 begins an uncontrolled increase. If not stopped orotherwise controlled, this voltage increase will rapidly exceed theoperating limitations of switching transistor 14 and energy storagecapacitor 56 and result in serious damage to the power supply circuit.

In order to solve this circuit design problem, the FIG. 1 circuitincorporates an overvoltage protection circuit which includes outputvoltage sensing means consisting of series coupled semiconductor means72 and biasing means 74.

Semiconductor means 72 in the preferred embodiment of the inventiontakes the form of two series coupled sidacs 76 and 78. Sidac 76 iscoupled to voltage conductor 70 by voltage scaling means with a one toone voltage ratio in the form of a direct electrical connection. Thespecific sidacs selected for use in the FIG. 1 circuit each have a onehundred and thirty volt breakover voltage. As illustrated in FIG. 5A,the current flow through a sidac semiconductor device remains at anear-zero level until the sidac voltage reaches the breakover voltage.At the breakover voltage, the sidac transitions from a normal resistivemode of operation into a negative resistance mode of operationdesignated the foldback region in FIG. 5A.

In the preferred embodiment of the FIG. 1 converter, the overvoltageprotection circuit is designed to limit the maximum output voltage totwo hundred and sixty volts. As the voltage level on voltage conductor70 begins to exceed two hundred and sixty volts, sidacs 76 and 78 areoperating at their breakover voltage where they begin to conduct currentand enter the negative resistance foldback region. The flow of currentthrough sidacs 76 and 78 is transmitted to resistive biasing means 74which results in the generating of an overvoltage signal which iscoupled to the base of base drive shunt transistor 24 of the powersupply disabling means. This overvoltage signal activates transistor 24and shunts all base drive current away from switching transistor 14 toground and causes switching transistor 14 to transition from theconductive state into the non-conductive state. With switchingtransistor 14 in the non-conductive state, the power supply outputvoltage on voltage conductor 70 ramps downward for a period of time.After a comparatively short but measurable time, the voltage impressedacross sidacs 76 and 78 drops to a level below the lowest voltagecapable of maintaining the sidacs in the foldback region and the sidacsthen transition from operation in their negative resistance region intothe normal resistive mode of operation. When this transition occurs, theflow of current through sidacs 76 and 78 terminates, resulting in theremoval of the overvoltage signal from the base of transistor 24. Basedrive current is once again directed to the base terminal of switchingtransistor 14, causing it to resume normal operation.

If the circuit defect which initially activated the overvoltageprotection circuit has not been cured, the resumed converter operationwill once again begin increasing the voltage on energy storage capacitor56. When the maximum desired output voltage is achieved on voltageconductor 70, the overvoltage protection circuit is activated onceagain. This cycle of activating, deactivating and reactivating theovervoltage protection circuit causes the output voltage on voltageconductor 70 to fluctuate from the maximum desired output voltage to avoltage level somewhat below that level until the circuit fault isultimately cured. Because this circuit prevents overvoltage conditions,circuit defects of the type discussed above cannot damage any elementsof the FIG. 1 power supply circuit.

A zener diode which completely breaks down and commences full currentconduction at a predetermined breakdown voltage cannot successfully beused as a substitute for the semiconductor means of the presentovervoltage protection circuit. Substituting a zener diode for sidacs 76and 78 would merely result in circuit oscillation about the maximumdesired output voltage and would not serve the purposes of the presentinvention.

Referring now to 5B, the voltage versus current plot of a relatedsemiconductor device designated a diac is illustrated. Although sidacsare preferred over diacs, a diac possesses the required breakovervoltage/foldback region characteristics necessary for proper operationof the overvoltage protection circuit of the present invention.

FIG. 5C illustrates the voltage versus current characteristics ofanother semiconductor device referred to as a silicon bilateral switch(SBS) which also possesses a breakover voltage/foldback regioncharacteristic of the type necessary for proper operation of theovervoltage protection circuit of the present invention. Semiconductormeans in the form of a sidac is strongly preferred over either the diacor SBS alternatives discussed above.

Although one terminal of sidac 76 is shown directly coupled to voltageconductor 70, that specific type of circuit configuration is notnecessary for proper operation of the overvoltage protection circuit. Aswould be immediately recognized by a person or ordinary skill in theart, a voltage scaling network or voltage divider network could becoupled across voltage conductor 70 and ground to scale the inputvoltage to a sidac to thereby render a sidac having a fixed breakovervoltage rating capable of functioning to activate the overvoltageprotection circuit at converter output voltages substantially differentthan the sidac breakover voltage. Although a pair of sidacs have beendepicted as being coupled in series in FIG. 1, it would be readilyapparent that a single sidac may be used alone, that a variable numberof sidacs could be coupled in series or that one or more sidacs could beused in connection with various voltage scaling means to cause anovervoltage protection circuit of the present invention to commenceoperation at any predetermined voltage.

FIG. 2 illustrates another configuration of variable impedance means 52in the output circuit of the DC to DC converter illustrated in FIG. 1.

It will be apparent to those skilled in the art that the disclosed drivecurrent regulator circuit, afterglow prevention circuit and overvoltageprotection circuit for a DC to DC converter may be modified in numerousways and may assume many embodiments other than the preferredembodiments specifically set out and described above. For example, theFIG. 1 circuit has been discussed in connection with a twelve toforty-eight volts DC input voltage capacity although it is readilyevident that the circuit elements could easily be modified to functionnormally with numerous other input voltage ranges. Accordingly, it isintended by the appended claims to cover all such modifications of theinvention which fall within the true spirit and scope of the invention.

I claim:
 1. Apparatus for delivering current to an intermittentlyenergized gaseous discharge tube having a low impedance conductive stateand a high impedance non-conductive state, comprising:a. a DC to DCconverter for receiving a DC input voltage on a converter inputterminal, for generating a pulsed output voltage on a converter outputterminal and for periodically energizing the gaseous discharge tube intothe low impedance conductive state, the apparatus defining a firstcurrent flow path for enabling the DC input voltage to cause a DCcurrent to flow through the converter to the gaseous discharge tube; b.energy storage means coupled between the converter output terminal andthe gaseous discharge tube for periodically receiving and storing energyfrom the converter when the gaseous discharge tube is in thenon-conductive state and for periodically transferring energy into thegaseous discharge tube when the tube is in the conductive state, theapparatus defining a second current flow path from the converter outputterminal to the energy storage means and a third current flow path fromthe energy storage means to the gaseous discharge tube; and c. a currentlimiting device coupled in series with the first current flow pathbetween the converter output terminal and the energy storage means forlimiting the flow of DC current from the converter input terminal intothe gaseous discharge tube to thereby prevent afterglow of the gaseousdischarge tube caused by feedthrough of the DC input voltage.
 2. Theapparatus of claim 1 wherein said current limiting device prevents theflow of DC current from the converter input terminal into the gaseousdischarge tube.
 3. The apparatus of claim 2 wherein the current limitingdevice includes a capacitor.
 4. The apparatus of claim 1 wherein the DCto DC converter includes a variable frequency converter and wherein thefrequency of the pulsed output voltage of the converter varies between alow operating frequency and a high operating frequency in response tovariations in the output voltage of the converter between a low outputvoltage and a high output voltage.
 5. The apparatus of claim 4 whereinthe current limiting device includes a capacitor and wherein the valueof the capacitor is selected to present a low impedance at the highconverter operating frequency and a high impedance at the low converteroperating frequency.
 6. The apparatus of claim 5 wherein the operatingfrequency of the converter varies between a low operating frequency offrom about 5000 Hz to about 10,000 Hz to a high operating frequency offrom about 40,000 Hz to about 50,000 Hz.
 7. The apparatus of claim 1wherein the converter includesa. a coupled inductor having a primarywinding with a first terminal coupled to the converter input terminaland a second terminal; b. a switching transistor having a collectorcoupled to the second terminal of the primary winding; and c. thecurrent limiting device having a first terminal coupled to the secondterminal of the primary winding of the coupled inductor and a secondterminal coupled to the energy storage capacitor and to the gaseousdischarge tube.
 8. The apparatus of claim 7 wherein the current limitingdevice includes a capacitor.
 9. The apparatus of claim 1 wherein the DCto DC converter runs continuously without being periodically disabled.10. The apparatus of claim 9 wherein said current limiting devicerestricts the flow of DC current from the converter input terminal intothe gaseous discharge tube.
 11. The apparatus of claim 10 wherein thecurrent limiting device includes a capacitor.
 12. The apparatus of claim10 wherein the DC to DC converter includes a variable frequencyconverter and wherein the frequency of the pulsed output voltage of theconverter varies between a low operating frequency and a high operatingfrequency in response to variations in the output voltage of theconverter between a low output voltage and a high output voltage. 13.The apparatus of claim 12 wherein the current limiting device includes acapacitor and wherein the value of the capacitor is selected to presenta low impedance at the high converter operating frequency and a highimpedance at the low converter operating frequency.
 14. The apparatus ofclaim 13 wherein the operating frequency of the converter varies betweena low operating frequency of from about 5000 Hz to about 10,000 Hz to ahigh operating frequency of from about 40,000 Hz to about 50,000 Hz. 15.The apparatus of claim 10 wherein the converter includesa. a coupledinductor having a primary winding with a first terminal coupled to theconverter input terminal and a second terminal; b. a switchingtransistor having a collector coupled to the second terminal of theprimary winding; and c. the current limiting device having a firstterminal coupled to the second terminal of the primary winding of thecoupled inductor and a second terminal coupled to the energy storagecapacitor and to the gaseous discharge tube.
 16. The apparatus of claim1 wherein the current limiting device includes a capacitor.